This invention relates to rectifier for boost DC-DC converters. More particularly, this invention relates to synchronous rectifiers for boost DC-DC converters that can be fabricated in conventional bipolar processes.
A boost DC-DC converter generates a output voltage higher than its input voltage. Convertional boost DC-DC converters (also known as regulators) typically employ an inductive energy storage lement (e.g., an inductor) coupled in series with a "free-wheeling" diode rectifier. The diode rectifier conducts current only in the direction of the load, and is considered free-wheeling because its operation cannot be controlled independent of the voltages at its anode and cathode (i.e., conduction occurs when voltage at the anode exceeds voltage at the cathode by the turn-on voltage, which is typically 0.3 to 0.7 volts).
Boost DC-DC converters typically include a switch that provides a current path to ground from a node formed between the inductive energy storage element and the diode rectifier. Control circuitry regulates the duty cycle of the switch (i.e., the percentage of time the switch is ON during a cycle of operation). The duty cycle controls the amount of energy delivered to the load.
FIG. 1 shows a known conventional boost DC-DC converter, which can be fabricated in a conventional bipolar process. Such a process produces high performance NPN transistors that have high current gain and good current-carrying capabilities. Furt lermore, lateral and substrate PNP transistors can also be fabricated without additional processing steps (although they are generally slower than NPN transistors and have limited current-carrying capabilities). In contrast, other known boost converters (three of which are described belo ) are fabricated in BICMOS or complementary bipolar processes. These processes require additional processing steps, and are thus more complex and typically more costly than conventional bipolar processes.
Converter 100 includes inductor 102, power NPN transistor switch 104, and diode rectifier 10. The duty cycle of switch 104 is controlled by switch control circuitry 108. During switch 104 ON-time (i.e., when switch 104 is conducting), current flows from input voltage source VIN through inductor 102 and switch 104 to ground. The voltage across inductor 102 is positive (V.sub.L .apprxeq.V.sub.IN &gt;0). This positive voltage causes magnetic flux in inductor 102 to increase. Because magnetic flux is proportional to inductor current, inductor current increases when magnetic flux increases.
During switch 104 OFF-time (i.e., when switch 104 is non-conducting), current flows from V.sub.IN through inductor 102 and diode 106 to output 120. At steady-state (i.e., during normal boost operation, also known as step-up mode), output voltage VOUT is higher than V.sub.IN. The voltage across inductor 102 is therefore negative (V.sub.L =V.sub.IN -V.sub.DIODE -V.sub.OUT &lt;0, where V.sub.DIODE is the voltage drop across diode 106). This negative voltage causes magnetic flux in inductor 102 to decrease. Accordingly, inductor current decreases. Preferably, the duty cycle of switch 104 is set such that flux increases during ON-time equal flux decreases during OFF-time. The net DC flux in inductor 102 is therefore zero, permitting inductor 102 to establish volt-second balance (i.e., equilibrium).
A disadvantage of converter 100 occurs during converter start-up when the output voltage rises from zero to its steady-state value. During switch 104 ON-time, inductor voltage is positive and magnetic flux increases as in steady-state. But, during switch 104 OFF-time, inductor voltage remains positive (V.sub.L =V.sub.IN -V.sub.DIODE -V.sub.OUT &gt;0) while V.sub.OUT is less than V.sub.IN by at least V.sub.DIODE. Inductor flux therefore continues to increase. Moreover, because the output bypass capacitance (represented by capacitor 106) is typically high (e.g., 100-200 .mu.F), the rising output voltage transient is slow. Inductor current therefore continues to increase until capacitor 160 is charged to V.sub.IN -V.sub.DIODE. This results in high "in-rush" current from inductor 102.
High in-rush current causes excessive power dissipation in switch 104 and diode 106, and can damage some types of batteries commonly used as V.sub.IN by exceeding their maximum current rating. Furthermore, high in-rush current cannot be prevented by modifying the duty cycle of switch 104, because inductor voltage is positive during both switch 104 ON-time and OFF-time, rendering ineffective any duty cycle modifications.
Another disadvantage of converter 100 is that the output cannot be electrically isolated from the input (i.e., converter 100 does not have a shutdown mode). Current can always flow from input 124 through inductor 102 and diode 106 to output 120. Converter 100 would need an extra switch coupled in series with inductor 102 and diode 106 to effectively disconnect the output from the input. A shutdown mode isolates the load during converter shutdown to prevent unnecessary power dissipation.
FIG. 2 shows a known boost DC-DC converter with a PMOS synchronous rectifier. A synchronous rectifier is a controllable switch that replaces the rectifying diode. The synchronous rectifier turns ON substantially immediately after the switch turns OFF. Converter 200 includes inductor 202, power NMOS transistor switch 204, which is driven by switch control circuitry 208, PMOS transistor synchronous rectifier 206, which is driven by rectifier control circuitry 218, and PMOS shutdown transistor 214, which is driven by shutdown control circuitry 216. Diodes 210 and 212 represent parasitic body diodes of PMOS rectifier 20. Converter 200 may be, for example, the ML4875 of Micro Linear Corporation, of San Jose, Calif.
Unlike converter 100, converter 200 includes a shutdown mode. When converter 200 receives a shutdown signal, shutdown control circuitry 216 causes switch control circuitry 208 and rectifier control circuitry 218 to respectively turn OFF switch 204 and rectifier 206 (inputs to switch control circuitry 208 and rectifier control circuitry 218 are not shown in FIG. 2 for clarity). Shutdown control circuitry 216 also turns OFF transistor 214, which is ON during normal boost operation. With rectifier 206 and transistor 214 both OFF, a current path from inductor 202 to output 220 no longer exists. Output 220 is thus effectively disconnected from input 224. Output voltage V.sub.OUT then decreases as output capacitor 260 discharges through a load (not shown) coupled to output 220.
Because of efficiency concerns, switch 204 and rectifier 206 are preferably not ON simultaneously. During the interval between switch 204 turning OFF and rectifier 206 turning ON, inductor current is carried by free-wheeling diode 210 and transistor 214 (which shorts diode 212). Transistor 214 therefore carries as much current as switch 204 and rectifier 206, and is accordingly as large. Thus, by implementing a shutdown mode with a separate shutdown transistor, additional circuit die space is required to implement converter 200.
In normal boost mode, converter 200 operates similarly to converter 100. During switch 204 ON-time, rectifier 206 is OFF and voltage across inductor 202 is positive. During switch 204 OFF-time, rectifier 206 is ON, delivering inductor current to output 220, and voltage across inductor 202 is negative. Inductor 202 can therefore establish volt-second balance.
However, during converter start-up, converter 200 has the same disadvantage as converter 100--converter 200 cannot prevent the development of high in-rush current. While V.sub.OUT is less than V.sub.IN by at least the voltage drop across diode 210, inductor voltage remains positive during switch 204 OFF-time regardless of the bias on the gate of PMOS rectifier 206, because diode 210 and transistor 214 provide a parallel current path to output 220. Inductor 202 therefore cannot establish volt-second balance, which can result in high in-rush current.
Another disadvantage of converter 200 is that it is typically fabricated in a BICMOS process in which PMOS transistors are fabricated in an N-epitaxial layer and NMOS transistors are fabricated in P-wells tied to the P-substrate. Such BICMOS processes are more complex and typically more costly than conventional bipolar or conventional CMOS processes. In particular, for example, converter 200 requires an N-buried layer under PMOS transistor 214 to limit substrate current through diode 210. Conventional CMOS processes typically do not have buried layers. In the absence of an N-buried layer, conduction by diode 210, which occurs during the interval after switch 204 turns OFF and before PMOS rectifier 206 turns ON, results in current flow into the substrate. If diode 210 current is high, substrate current will be substantial. This can result in the unintentional forward biasing of P-substrate and N-epitaxial layer junctions, which will adversely affect the operation of converter 200. Thus, converter 200 cannot likely be fabricated in a conventional process.
FIG. 3 shows a known boost/step-down DC-DC converter with a power PNP synchronous rectifier. Converter 300 controls in-rush current, includes a shutdown mode, and also operates in step-down mode (i.e., the steady-state output voltage is less than the input voltage). Converter 300 includes inductor 302, NPN transistor switch 304, and PNP transistor synchronous rectifier 306. Switch control circuitry 308 drives switch 304 and rectifier control circuitry 318 drives rectifier 306. Such a converter may be, for example, a MAX877L/878L/879L boost/step-down regulator of Maxim Integrated Products, of Sunnyvale, Calif.
Converter 300 limits the development of high in-rush current. During switch 304 ON-time, voltage across inductor 302 is positive (V.sub.L =V.sub.IN &gt;0). During switch 304 OFF-time, the base of PNP rectifier 306 is driven by rectifier control circuitry 318 such that the voltage at node 322 (V.sub.322 V.sub.EB306 +V.sub.BASE306) is higher than V.sub.IN, regardless of the output voltage. The voltage across inductor 302 is therefore negative while PNP rectifier 306 delivers inductor current to output 320, which is coupled to a load (not shown). Inductor 302 can thus establish volt-second balance, limiting the development of high in-rush current.
In shutdown mode, rectifier control circuitry 318 turns PNP transistor 306 OFF, effectively disconnecting output 320 from input 324. Output voltage V.sub.OUT then decreases as output capacitor 360 discharges through the load.
Converter 300 however is typically fabricated in a complementary bipolar process. Although such processes produce vertical PNP transistors that have performance and current-carrying capabilities comparable to those of NPN transistors, complementary bipolar processes are more complex and typically more costly than conventional bipolar processes. Moreover, converter 300 cannot easily be fabricated in a conventional bipolar process. For example, PNP transistor 306, which carries a substantial amount of inductor current, would have to be fabricated as an excessively large lateral PNP transistor to carry the same amount of inductor current. Such a large PNP transistor would require additional circuit die space, which is typically limited and costly in most converter designs.
FIG. 4 shows a known boost DC-DC converter with an NMOS synchronous rectifier. Converter 400 includes a shutdown mode and an auxiliary boost output, and is typically fabricated in a BICMOS process. Converter 400 includes inductor 402, NMOS transistor synchronous rectifier 406, NMOS transistor switch 404 auxiliary boost output VGD, and variable frequency control circuitry 418. Diodes 410 and 412 represent the parasitic body diodes of rectifier 406. Converter 400 may be, for example, a UCC3941 boost converter of Unitrode Corporation, of Merrimack, N.H.
Converter 400 operates efficiently when the gate drive voltage of NMOS rectifier 406 is higher than V.sub.OUT by about 1-2 volts. The gate drive voltage is supplied by auxiliary boost output V.sub.GD, which typically has a maximum voltage of about 8 volts. The power transfer from input 424 is multiplexed between V.sub.OUT and V.sub.GD. Before V.sub.GD reaches 7.5 volts, rectifier 406 remains OFF and converter 400 supplies V.sub.GD. Once V.sub.GD increases above 7.5 volts, V.sub.OUT has priority in the multiplexing scheme. In shutdown mode, rectifier 406 is turned OFF and body diode 410 blocks current flow from input 424 to output 420, effectively disconnecting input 424 from output 420.
A disadvantage of converter 400 is that the gate threshold voltage of rectifier 406 increases as its source voltage increases. This is known as the body effect. It results from the reverse-bias between the source of rectifier 406 (which is coupled to output 420) and its body at node 411. If node 411 could be tied to output 420, the body effect can be eliminated. Such a connection however is not likely in this BICMOS process. Therefore, because the source of rectifier 406 is coupled to output 420, the gate threshold voltage of rectifier 406 increases when V.sub.OUT increases. Thus, if V.sub.OUT causes the gate threshold voltage to increase to a level where V.sub.GD can no longer supply a gate drive voltage of about 1-2 volts above V.sub.OUT, the efficiency of converter 400 decreases.
Another disadvantage of converter 400 occurs during converter start-up if rectifier 406 is driven into the triode region (also known as the ohmic or nonsaturation region), where V.sub.DS is low. Low V.sub.DS results in positive inductor voltage while V.sub.OUT is less than V.sub.IN, preventing inductor 402 from establishing volt-second balance. This can result in high in-rush current.
In view of the foregoing, it would be desirable to be able to provide a boost DC-DC converter that limits inductive in-rush current.
It would also be desirable to be able to provide a boost DC-DC converter that includes a shutdown mode.
It would further be desirable to be able to provide a boost DC-DC converter that also operates in step-down mode.
It would still further be desirable to be able to provide a boost DC-DC converter that can be fabricated in a conventional bipolar process.